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FEATURES
s s s s s s s s s s s s s s s
LTC3728L/LTC3728LX Dual, 550kHz, 2-Phase Synchronous Regulators DESCRIPTIO
The LTC(R)3728L/LTC3728LX are dual high performance step-down switching regulator controllers that drive all N-channel synchronous power MOSFET stages. A constant frequency current mode architecture allows phaselockable frequency of up to 550kHz. Power loss and noise due to the ESR of the input capacitors are minimized by operating the two controller output stages out of phase. OPTI-LOOP compensation allows the transient response to be optimized over a wide range of output capacitance and ESR values. The precision 0.8V reference and power good output indicator are compatible with future microprocessor generations, and a wide 4.5V to 28V (30V maximum) input supply range encompasses all battery chemistries. A RUN/SS pin for each controller provides both soft-start and optional timed, short-circuit shutdown. Current foldback limits MOSFET dissipation during short-circuit conditions when overcurrent latchoff is disabled. Output overvoltage protection circuitry latches on the bottom MOSFET until VOUT returns to normal. The FCB mode pin can select among Burst Mode, constant frequency mode and continuous inductor current mode or regulate a secondary winding. The LTC3728L/LTC3728LX include a power good output pin that indicates when both outputs are within 7.5% of their designed set point.
, LTC and LT are registered trademarks of Linear Technology Corporation. Burst Mode and OPTI-LOOP are registered trademarks of Linear Technology Corporation.
s
Dual, 180 Phased Controllers Reduce Required Input Capacitance and Power Supply Induced Noise OPTI-LOOP(R) Compensation Minimizes COUT 1% Output Voltage Accuracy (LTC3728LC) Power Good Output Voltage Indicator Phase-Lockable Fixed Frequency 250kHz to 550kHz Dual N-Channel MOSFET Synchronous Drive Wide VIN Range: 4.5V to 28V Operation Very Low Dropout Operation: 99% Duty Cycle Adjustable Soft-Start Current Ramping Foldback Output Current Limiting Latched Short-Circuit Shutdown with Defeat Option Output Overvoltage Protection Low Shutdown IQ: 20A 5V and 3.3V Standby Regulators 3 Selectable Operating Modes: Constant Frequency, Burst Mode(R) Operation and PWM 5mm x 5mm QFN and 28-Lead Narrow SSOP Packages
APPLICATIO S
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Notebook and Palmtop Computers Telecom Systems Portable Instruments Battery-Operated Digital Devices DC Power Distribution Systems
TYPICAL APPLICATIO
+
4.7F M1 L1 3.2H D3 VIN PGOOD INTVCC TG1 CB1, 0.1F BOOST1 SW1 BG1 fIN 500kHz PLLIN SENSE1+ RSENSE1 0.01 VOUT1 5V 5A 1000pF SENSE1- VOSENSE1 CC1 220pF RC1 15k ITH1 CSS1 0.1F SENSE2 - VOSENSE2 ITH2 CSS2 0.1F CC2 220pF RC2 15k LTC3728L/ LTC3728LX TG2 BOOST2 SW2 BG2 PGND SENSE2 + 1000pF CB2, 0.1F D4
1F CERAMIC M2
+
R2 105k COUT1 1% 47F 6V SP M1, M2: FDS6982S
R1 20k 1%
RUN/SS1 SGND RUN/SS2
Figure 1. High Efficiency Dual 5V/3.3V Step-Down Converter
U
U
U
VIN 5.2V TO 28V CIN 22F 50V CERAMIC L2 3.2H
RSENSE2 0.01 VOUT2 3.3V 5A
R3 20k 1%
R4 63.4k 1%
COUT 56F 6V SP
+
3728 F01
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LTC3728L/LTC3728LX
ABSOLUTE
AXI U
RATI GS
Input Supply Voltage (VIN)........................ 30V to - 0.3V Top Side Driver Voltages (BOOST1, BOOST2) .................................. 36V to - 0.3V Switch Voltage (SW1, SW2) ........................ 30V to - 5V INTVCC, EXTVCC, RUN/SS1, RUN/SS2, (BOOST1-SW1), (BOOST2-SW2), PGOOD ............................ 7V to - 0.3V SENSE1+, SENSE2 +, SENSE1-, SENSE2 - Voltages ....................... (1.1)INTVCC to - 0.3V PLLIN, PLLFLTR, FCB, Voltage ........... INTVCC to - 0.3V
PACKAGE/ORDER I FOR ATIO
TOP VIEW RUN/SS1 SENSE1+ SENSE- VOSENSE1 PLLFLTR PLLIN FCB ITH1 SGND 1 2 3 4 5 6 7 8 9 28 PGOOD 27 TG1 26 SW1 25 BOOST1 24 VIN 23 BG1 22 EXTVCC 21 INTVCC 20 PGND 19 BG2 18 BOOST2 17 SW2 16 TG2 15 RUN/SS2
RUN/SS1
SENSE1-
SENSE1+
PGOOD
LTC3728LCGN LTC3728LEGN
32 31 30 29 28 27 26 25 VOSENSE1 1 PLLFLTR 2 PLLIN 3 FCB 4 ITH1 5 SGND 6 3.3VOUT 7 ITH2 8 9 10 11 12 13 14 15 16 33 24 BOOST1 23 VIN 22 BG1 21 EXTVCC 20 INTVCC 19 PGND 18 BG2 17 BOOST2
SW1
TG1
NC
NC
ORDER PART NUMBER
3.3VOUT 10 ITH2 11 VOSENSE2 12 SENSE2- 13 SENSE2+ 14
SENSE2-
VOSENSE2
SENSE2+
NC
TG2
RUN/SS2
SW2
NC
GN PACKAGE 28-LEAD NARROW PLASTIC SSOP
TJMAX = 125C, JA = 95C/W
Consult LTC Marketing for parts specified with wider operating temperature ranges.
The q denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25C. VIN = 15V, VRUN/SS1, 2 = 5V unless otherwise noted.
SYMBOL VOSENSE1, 2 IVOSENSE1, 2 VREFLNREG VLOADREG PARAMETER Regulated Feedback Voltage Feedback Current CONDITIONS (Note 3); ITH1, 2 Voltage = 1.2V (LTC3728LC) q (Note 3); ITH1, 2 Voltage = 1.2V (LTC3728LE/LTC3728LX) q (Note 3) MIN 0.792 0.788 TYP 0.800 0.800 -5 0.002
q q
ELECTRICAL CHARACTERISTICS
Main Control Loops 0.808 0.812 - 50 0.02 0.5 - 0.5 V V nA %/V % % mmho
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Reference Voltage Line Regulation VIN = 3.6V to 30V (Note 3) Output Voltage Load Regulation (Note 3) Measured in Servo Loop; ITH Voltage = 1.2V to 0.7V Measured in Servo Loop; ITH Voltage = 1.2V to 2.0V Transconductance Amplifier gm ITH1, 2 = 1.2V; Sink/Source 5uA; (Note 3)
gm1, 2
2
U
U
W
WW U
W
(Note 1)
ITH1, ITH2, VOSENSE1, VOSENSE2 Voltages ...2.7V to - 0.3V Peak Output Current <10s (TG1, TG2, BG1, BG2) .. 3A INTVCC Peak Output Current ................................ 40mA Operating Temperature Range (Note 7) LTC3728LC/LTC3728LXC ....................... 0C to 85C LTC3728LE ........................................ - 40C to 85C Junction Temperature (Note 2) ............................ 125C Storage Temperature Range ................ - 65C to 125C Reflow Peak Body Temperature (UH Package) .... 260C Lead Temperature (Soldering, 10 sec) (GN Package) ................................................... 300C
TOP VIEW
ORDER PART NUMBER LTC3728LCUH LTC3728LEUH LTC3728LXCUH
UH PART MARKING 3728L 3728LE 3728LX
UH PACKAGE 32-LEAD (5mm x 5mm) PLASTIC QFN
TJMAX = 125C, JA = 34C/W EXPOSED PAD IS SGND (PIN 33), MUST BE SOLDERED TO PCB
MAX
UNITS
0.1 - 0.1 1.3
LTC3728L/LTC3728LX
The q denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25C. VIN = 15V, VRUN/SS1, 2 = 5V unless otherwise noted.
SYMBOL gmGBW1, 2 IQ PARAMETER Transconductance Amplifier GBW Input DC Supply Current Normal Mode Shutdown Forced Continuous Threshold Forced Continuous Pin Current Burst Inhibit (Constant Frequency) Threshold Undervoltage Lockout Feedback Overvoltage Lockout Sense Pins Total Source Current Maximum Duty Factor Soft-Start Charge Current VFCB = 0.85V Measured at FCB pin VIN Ramping Down Measured at VOSENSE1, 2 (Each Channel); VSENSE1-, 2 - = VSENSE1+, 2+ = 0V In Dropout VRUN/SS1, 2 = 1.9V VRUN/SS1, VRUN/SS2 Rising Soft Short Condition VOSENSE1, 2 = 0.5V; VRUN/SS1, 2 = 4.5V VOSENSE1, 2 = 0.5V VOSENSE1, 2 = 0.7V,VSENSE1-, 2 - = 5V VOSENSE1, 2 = 0.7V,VSENSE1-, 2 - = 5V (Note 5) CLOAD = 3300pF CLOAD = 3300pF (Note 5) CLOAD = 3300pF CLOAD = 3300pF CLOAD = 3300pF Each Driver CLOAD = 3300pF Each Driver Tested with a Square Wave (Note 6) 6V < VIN < 30V, VEXTVCC = 4V ICC = 0 to 20mA, VEXTVCC = 4V ICC = 20mA, VEXTVCC = 5V ICC = 20mA, EXTVCC Ramping Positive
q q q q
ELECTRICAL CHARACTERISTICS
CONDITIONS ITH1, 2 = 1.2V; (Note 3) (Note 4) VIN = 15V; EXTVCC Tied to VOUT1; VOUT1 = 5V VRUN/SS1, 2 = 0V
q
MIN
TYP 3 450 20
MAX
UNITS MHz A A V A V V V A % A
35 0.84 - 0.1 4.8 4 0.88
VFCB IFCB VBINHIBIT UVLO VOVL ISENSE DFMAX IRUN/SS1, 2
0.76 - 0.50
0.800 - 0.18 4.3 3.5
0.84 - 90 98 0.5 1.0 0.5
0.86 - 60 99.4 1.2 1.5 4.1 2 1.6
VRUN/SS1, 2 ON RUN/SS Pin ON Threshold ISCL1, 2 ISDLHO VSENSE(MAX) RUN/SS Discharge Current Shutdown Latch Disable Current Maximum Current Sense Threshold TG Transition Time: Rise Time Fall Time BG Transition Time: Rise Time Fall Time Top Gate Off to Bottom Gate On Delay Synchronous Switch-On Delay Time Bottom Gate Off to Top Gate On Delay Top Switch-On Delay Time Minimum On-Time Internal VCC Voltage INTVCC Load Regulation EXTVCC Voltage Drop EXTVCC Switchover Voltage EXTVCC Hysteresis Nominal Frequency Lowest Frequency Highest Frequency PLLIN Input Resistance Phase Detector Output Current Sinking Capability Sourcing Capability
2.0 4.75 4 5 85 88 100 100 100 90
V V A A mV mV ns ns ns ns ns ns ns
VRUN/SS1, 2 LT RUN/SS Pin Latchoff Arming Threshold VRUN/SS1, VRUN/SS2 Rising from 3V
65 62
75 75 55 55 45 45 80 80 100
TG1, 2 tr TG1, 2 tf BG1, 2 tr BG1, 2 tf TG/BG t1D BG/TG t2D tON(MIN) VINTVCC VLDO INT VLDO EXT VEXTVCC VLDOHYS fNOM fLOW fHIGH RPLLIN I PLLFLTR
INTVCC Linear Regulator 4.8 5.0 0.2 100 4.5 4.7 0.2 VPLLFLTR = 1.2V VPLLFLTR = 0V VPLLFLTR 2.4V 360 230 480 400 260 550 50 fPLLIN < fOSC fPLLIN > fOSC -15 15 440 290 590 5.2 2.0 200 V % mV V V kHz kHz kHz k A A
Oscillator and Phase-Locked Loop
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LTC3728L/LTC3728LX
The q denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25C. VIN = 15V, VRUN/SS1, 2 = 5V unless otherwise noted.
SYMBOL V3.3OUT V3.3IL V3.3VL PGOOD Output VPGL IPGOOD VPG PGOOD Voltage Low PGOOD Leakage Current PGOOD Trip Level, Either Controller IPGOOD = 2mA VPGOOD = 5V VOSENSE with Respect to Set Output Voltage VOSENSE Ramping Negative VOSENSE Ramping Positive -6 6 -7.5 7.5 0.1 0.3 1 - 9.5 9.5 V A % % PARAMETER 3.3V Regulator Output Voltage 3.3V Regulator Load Regulation 3.3V Regulator Line Regulation CONDITIONS No Load I3.3 = 0 to 10mA 6V < VIN < 30V
q
ELECTRICAL CHARACTERISTICS
3.3V Linear Regulator
MIN 3.2
TYP 3.35 0.5 0.05
MAX 3.45 2 0.2
UNITS V % %
Note 1: Absolute Maximum Ratings are those values beyond which the life of a device may be impaired. Note 2: TJ is calculated from the ambient temperature TA and power dissipation PD according to the following formulas: LTC3728LUH/LTC3728LXUH: TJ = TA + (PD * 34C/W) LTC3728LGN: TJ = TA + (PD * 95C/W) Note 3: The IC is tested in a feedback loop that servos VITH1, 2 to a specified voltage and measures the resultant VOSENSE1, 2. Note 4: Dynamic supply current is higher due to the gate charge being
delivered at the switching frequency. See Applications Information. Note 5: Rise and fall times are measured using 10% and 90% levels. Delay times are measured using 50% levels. Note 6: The minimum on-time condition is specified for an inductor peak-to-peak ripple current 40% of IMAX (see minimum on-time considerations in the Applications Information section). Note 7: The LTC3728LC/LTC3728LXC are guaranteed to meet performance specifications from 0C to 85C. The LTC3728LE is guaranteed to meet performance specifications over the -40C to 85C operating temperature range as assured by design, characterization and correlation with statistical process controls.
TYPICAL PERFOR A CE CHARACTERISTICS
Efficiency vs Output Current and Mode (Figure 13)
100 90 80 Burst Mode OPERATION 100 VIN = 7V 90 90
EFFICIENCY (%)
EFFICIENCY (%)
60 50 40 30 20 10 0 0.001
80
EFFICIENCY (%)
70
FORCED CONTINUOUS MODE (PWM) CONSTANT FREQUENCY (BURST DISABLE) VIN = 15V VOUT = 5V f = 250kHz 0.1 0.01 1 OUTPUT CURRENT (A) 10
3728L G01
4
UW
Efficiency vs Output Current (Figure 13)
100
Efficiency vs Input Voltage (Figure 13)
VIN = 10V VIN = 15V VIN = 20V
80
70
70
60 VOUT = 5V f = 250kHz 50 0.001 0.1 0.01 1 OUTPUT CURRENT (A)
60
50 10
3728L G02
VOUT = 5V IOUT = 3A f = 250kHz 5 25 15 INPUT VOLTAGE (V) 35
3728L G03
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LTC3728L/LTC3728LX TYPICAL PERFOR A CE CHARACTERISTICS
Supply Current vs Input Voltage and Mode (Figure 13)
1000
200
INTVCC AND EXTVCC SWITCH VOLTAGE (V)
800
SUPPLY CURRENT (A)
EXTVCC VOLTAGE DROP (mV)
600 BOTH CONTROLLERS ON 400
200 SHUTDOWN 0 0 5 20 15 10 INPUT VOLTAGE (V) 25 30
3728L G04
Internal 5V LDO Line Regulation
5.1 5.0
ILOAD = 1mA
INTVCC VOLTAGE (V)
4.9 4.8 4.7 4.6 4.5 4.4 0 5 20 15 10 INPUT VOLTAGE (V) 25 30
3728L G07
VSENSE (mV)
VSENSE (mV)
Maximum Current Sense Threshold vs VRUN/SS (Soft-Start)
80 VSENSE(CM) = 1.6V
80
60 VSENSE (mV)
VSENSE (mV)
72
VSENSE (mV)
40
20
64
0 0 1 2 3 VRUN/SS (V)
3728L G10
4
UW
5 6
EXTVCC Voltage Drop
5.05 5.00 4.95 4.90 4.85 4.80 4.75
INTVCC and EXTVCC Switch Voltage vs Temperature
INTVCC VOLTAGE
150
100
50
EXTVCC SWITCHOVER THRESHOLD
0
0
10
20 CURRENT (mA)
30
40
3728L G05
4.70 - 50 - 25
50 25 75 0 TEMPERATURE (C)
100
125
3728L G06
Maximum Current Sense Threshold vs Duty Factor
75
Maximum Current Sense Threshold vs Percent of Nominal Output Voltage (Foldback)
80 70 60
50
50 40 30 20 10
25
0 0 20 40 60 DUTY FACTOR (%) 80 100
3728L G08
0
50 100 0 25 75 PERCENT ON NOMINAL OUTPUT VOLTAGE (%)
3728L G09
Maximum Current Sense Threshold vs Sense Common Mode Voltage
90 80 76 70 60 50 40 30 20 10 0 -10 -20 60 0 1 3 4 2 COMMON MODE VOLTAGE (V) 5
3728L G11
Current Sense Threshold vs ITH Voltage
68
-30
0
0.5
1
1.5 VITH (V)
2
2.5
3728L G12
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LTC3728L/LTC3728LX TYPICAL PERFOR A CE CHARACTERISTICS
Load Regulation
0.0
2.5
NORMALIZED VOUT (%)
-0.1
-0.2
VITH (V)
1.5
ISENSE (A)
-0.3
0.5
-0.4
0
1
3 2 LOAD CURRENT (A)
Maximum Current Sense Threshold vs Temperature
80
4
78 DROPOUT VOLTAGE (V)
RUN/SS CURRENT (A)
VSENSE (mV)
76
74
72
70 -50
-25
50 25 0 75 TEMPERATURE (C)
Soft-Start Up (Figure 13)
VOUT 5V/DIV VRUN/SS 5V/DIV IL 2A/DIV
IL 2A/DIV
VIN = 15V VOUT = 5V
5ms/DIV
6
UW
FCB = 0V VIN = 15V FIGURE 13 4
3728L G13
VITH vs VRUN/SS
VOSENSE = 0.7V
SENSE Pins Total Source Current
100
2.0
50
0
1.0
-50
0
5
0
1
2
3 VRUN/SS (V)
4
5
6
3728L G14
-100
0
2
4
6
3728L G15
VSENSE COMMON MODE VOLTAGE (V)
Dropout Voltage vs Output Current (Figure 14)
VOUT = 5V
RUN/SS Current vs Temperature
1.8 1.6
3
1.4 1.2 1.0 0.8 0.6 0.4 0.2
2 RSENSE = 0.015 1 RSENSE = 0.010 0
100
125
0
0.5
1.0 1.5 2.0 2.5 3.0 OUTPUT CURRENT (A)
3.5
4.0
0 -50
-25
0 25 50 75 TEMPERATURE (C)
100
125
3728L G17
3728L G18
3728L G25
Load Step (Figure 13)
Load Step (Figure 13)
VOUT 200mV/DIV
VOUT 200mV/DIV
IL 2A/DIV
3728L G19
VIN = 15V 20s/DIV VOUT = 5V VPLLFLTR = 0V LOAD STEP = 0A TO 3A Burst Mode OPERATION
3728L G20
20s/DIV VIN = 15V VOUT = 5V VPLLFLTR = 0V LOAD STEP = 0A TO 3A CONTINUOUS MODE
3728L G21
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LTC3728L/LTC3728LX TYPICAL PERFOR A CE CHARACTERISTICS
Input Source/Capacitor Instantaneous Current (Figure 13)
IIN 2A/DIV VIN 200mV/DIV VSW1 10V/DIV VSW2 10V/DIV IL 0.5A/DIV IL 0.5A/DIV VIN = 15V VOUT = 5V VPLLFLTR = 0V VFCB = OPEN IOUT = 20mA 10s/DIV
3728L G23
VIN = 15V 1s/DIV VOUT1 = 5V, VOUT2 = 3.3V VPLLFLTR = 0V IOUT5 = IOUT3.3 = 2A
Current Sense Pin Input Current vs Temperature
35
CURRENT SENSE INPUT CURRENT (A)
VOUT = 5V
EXTVCC SWITCH RESISTANCE ()
33
FREQUENCY (kHz)
31
29
27
25 -50 -25
50 25 0 75 TEMPERATURE (C)
Undervoltage Lockout vs Temperature
3.50 3.45 3.40 3.35 3.30 3.25 3.20 -50 -25
SHUTDOWN LATCH THRESHOLDS (V)
UNDERVOLTAGE LOCKOUT (V)
UW
100
3728L G26
Burst Mode Operation (Figure 13)
VOUT 20mV/DIV
Constant Frequency (Burst Inhibit) Operation (Figure 13)
VOUT 20mV/DIV
3728L G22
VIN = 15V VOUT = 5V VPLLFLTR = 0V VFCB = 5V IOUT = 20mA
2s/DIV
3728L G24
EXTVCC Switch Resistance vs Temperature
10
Oscillator Frequency vs Temperature
700 600 VPLLFLTR = 2.4V
8
500 400 300 200 100 VPLLFLTR = 1.2V
6
4
VPLLFLTR = 0V
2
125
0 -50 -25
50 25 0 75 TEMPERATURE (C)
100
125
0 - 50 - 25
50 25 75 0 TEMPERATURE (C)
100
125
3728L G27
3728L G28
Shutdown Latch Thresholds vs Temperature
4.5 4.0 3.5 3.0 2.5 2.0 1.5 1.0 0.5 0 -50 -25 0 25 50 75 TEMPERATURE (C) 100 125 LATCHOFF THRESHOLD LATCH ARMING
50 25 75 0 TEMPERATURE (C)
100
125
3728L G29
3728L G30
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LTC3728L/LTC3728LX
PI FU CTIO S
VOSENSE1, VOSENSE2: Error Amplifier Feedback Input. Receives the remotely-sensed feedback voltage for each controller from an external resistive divider across the output. PLLFLTR: Filter Connection for Phase-Locked Loop. Alternatively, this pin can be driven with an AC or DC voltage source to vary the frequency of the internal oscillator. PLLIN: External Synchronization Input to Phase Detector. This pin is internally terminated to SGND with 50k. The phase-locked loop will force the rising top gate signal of controller 1 to be synchronized with the rising edge of the PLLIN signal. FCB: Forced Continuous Control Input. This input acts on both controllers and is normally used to regulate a secondary winding. Pulling this pin below 0.8V will force continuous synchronous operation. ITH1, ITH2: Error Amplifier Output and Switching Regulator Compensation Point. Each associated channels' current comparator trip point increases with this control voltage. SGND: Small Signal Ground. Common to both controllers, this pin must be routed separately from high current grounds to the common (-) terminals of the COUT capacitors. 3.3VOUT: Lnear Regulator Output. Capable of supplying 10mA DC with peak currents as high as 50mA. NC: No Connect. SENSE2 -, SENSE1 -: The (-) Input to the Differential Current Comparators. SENSE2 +, SENSE1 +: The (+) Input to the Differential Current Comparators. The ITH pin voltage and controlled offsets between the SENSE- and SENSE+ pins in conjunction with RSENSE set the current trip threshold. RUN/SS2, RUN/SS1: Combination of soft-start, run control inputs and short-circuit detection timers. A capacitor to ground at each of these pins sets the ramp time to full output current. Forcing either of these pins back below 1.0V causes the IC to shut down the circuitry required for that particular controller. Latchoff overcurrent protection is also invoked via this pin as described in the Applications Information section. TG2, TG1: High Current Gate Drives for Top N-Channel MOSFETs. These are the outputs of floating drivers with a voltage swing equal to INTVCC - 0.5V superimposed on the switch node voltage SW. SW2, SW1: Switch Node Connections to Inductors. Voltage swing at these pins is from a Schottky diode (external) voltage drop below ground to VIN. BOOST2, BOOST1: Bootstrapped Supplies to the Top Side Floating Drivers. Capacitors are connected between the boost and switch pins and Schottky diodes are tied between the boost and INTVCC pins. Voltage swing at the boost pins is from INTVCC to (VIN + INTVCC). BG2, BG1: High Current Gate Drives for Bottom (Synchronous) N-Channel MOSFETs. Voltage swing at these pins is from ground to INTVCC. PGND: Driver Power Ground. Connects to the sources of bottom (synchronous) N-channel MOSFETs, anodes of the Schottky rectifiers and the (-) terminal(s) of CIN. INTVCC: Output of the Internal 5V Linear Low Dropout Regulator and the EXTVCC Switch. The driver and control circuits are powered from this voltage source. Must be decoupled to power ground with a minimum of 4.7F tantalum or other low ESR capacitor. EXTVCC: External Power Input to an Internal Switch Connected to INTVCC. This switch closes and supplies VCC power, bypassing the internal low dropout regulator, whenever EXTVCC is higher than 4.7V. See EXTVCC connection in Applications section. Do not exceed 7V on this pin. VIN: Main Supply Pin. A bypass capacitor should be tied between this pin and the signal ground pin. PGOOD: Open-Drain Logic Output. PGOOD is pulled to ground when the voltage on either VOSENSE pin is not within 7.5% of its set point. Exposed Pad (UH Package Only): Signal Ground. Must be soldered to the PCB, providing a local ground for the control components of the IC, and be tied to the PGND pin under the IC.
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U
U
U
LTC3728L/LTC3728LX
FU CTIO AL DIAGRA
PLLIN FIN 50k PLLFLTR RLP CLP CLK1 OSCILLATOR CLK2 - + PGOOD VOSENSE1 - + - + VOSENSE2 - INTVCC 0.18A R6 FCB + R5 - FCB 3V 4.5V - + BINH + 0.74V 0.74V 0.86V 0.86V PHASE DET
3.3VOUT
+ -
0.8V
VREF
VIN VIN 4.8V EXTVCC + - 5V LDO REG 1.2A
SHDN RST 4(VFB)
+
5V
INTVCC
SGND (UH PACKAGE PAD)
INTERNAL SUPPLY
OPERATIO
(Refer to Functional Diagram)
Main Control Loop The IC uses a constant frequency, current mode stepdown architecture with the two controller channels operating 180 degrees out of phase. During normal operation, each top MOSFET is turned on when the clock for that channel sets the RS latch, and turned off when the main current comparator, I1, resets the RS latch. The peak
W
INTVCC DUPLICATE FOR SECOND CONTROLLER CHANNEL BOOST DB VIN DROP OUT DET S R Q Q TOP BOT FCB SW SWITCH LOGIC BOT B SHDN INTVCC BG PGND TG CB D1
U
U
U
+
CIN
TOP ON
COUT
0.55V
+ -
+
RSENSE
VOUT
I1
+ -
-
++
3mV
-
- +
I2
INTVCC
+ 30k SENSE - 30k SENSE
0.86V 4(VFB) SLOPE COMP 45k
45k 2.4V - EA + OV + - 0.86V ITH RUN SOFT START RUN/SS CC VFB 0.80V VOSENSE R2
R1
CC2
RC
6V
CSS
3728 FD/F02
Figure 2
inductor current at which I1 resets the RS latch is controlled by the voltage on the ITH pin, which is the output of each error amplifier EA. The VOSENSE pin receives the voltage feedback signal, which is compared to the internal reference voltage by the EA. When the load current increases, it causes a slight decrease in VOSENSE relative to the 0.8V reference, which in turn causes the ITH voltage to
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LTC3728L/LTC3728LX
OPERATIO
increase until the average inductor current matches the new load current. After the top MOSFET has turned off, the bottom MOSFET is turned on until either the inductor current starts to reverse, as indicated by current comparator I2, or the beginning of the next cycle. The top MOSFET drivers are biased from floating bootstrap capacitor CB, which normally is recharged during each off cycle through an external diode when the top MOSFET turns off. As VIN decreases to a voltage close to VOUT, the loop may enter dropout and attempt to turn on the top MOSFET continuously. The dropout detector detects this and forces the top MOSFET off for about 400ns every tenth cycle to allow CB to recharge. The main control loop is shut down by pulling the RUN/SS pin low. Releasing RUN/SS allows an internal 1.2A current source to charge soft-start capacitor CSS. When CSS reaches 1.5V, the main control loop is enabled with the ITH voltage clamped at approximately 30% of its maximum value. As CSS continues to charge, the ITH pin voltage is gradually released allowing normal, full-current operation. When both RUN/SS1 and RUN/SS2 are low, all controller functions are shut down, including the 5V and 3.3V regulators. Low Current Operation The FCB pin is a multifunction pin providing two functions: 1) to provide regulation for a secondary winding by temporarily forcing continuous PWM operation on both controllers; and 2) to select between two modes of low current operation. When the FCB pin voltage is below 0.8V, the controller forces continuous PWM current mode operation. In this mode, the top and bottom MOSFETs are alternately turned on to maintain the output voltage independent of direction of inductor current. When the FCB pin is below VINTVCC - 2V but greater than 0.8V, the controller enters Burst Mode operation. Burst Mode operation sets a minimum output current level before inhibiting the top switch and turns off the synchronous MOSFET(s) when the inductor current goes negative. This combination of requirements will, at low currents, force the ITH pin below a voltage threshold that will
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(Refer to Functional Diagram)
temporarily inhibit turn-on of both output MOSFETs until the output voltage drops. There is 60mV of hysteresis in the burst comparator B tied to the ITH pin. This hysteresis produces output signals to the MOSFETs that turn them on for several cycles, followed by a variable "sleep" interval depending upon the load current. The resultant output voltage ripple is held to a very small value by having the hysteretic comparator after the error amplifier gain block. Frequency Synchronization The phase-locked loop allows the internal oscillator to be synchronized to an external source via the PLLIN pin. The output of the phase detector at the PLLFLTR pin is also the DC frequency control input of the oscillator that operates over a 260kHz to 550kHz range corresponding to a DC voltage input from 0V to 2.4V. When locked, the PLL aligns the turn on of the top MOSFET to the rising edge of the synchronizing signal. When PLLIN is left open, the PLLFLTR pin goes low, forcing the oscillator to minimum frequency. Constant Frequency Operation When the FCB pin is tied to INTVCC, Burst Mode operation is disabled and the forced minimum output current requirement is removed. This provides constant frequency, discontinuous current (preventing reverse inductor current) operation over the widest possible output current range. This constant frequency operation is not as efficient as Burst Mode operation, but does provide a lower noise, constant frequency operating mode down to approximately 1% of the designed maximum output current. Continuous Current (PWM) Operation Tying the FCB pin to ground will force continuous current operation. This is the least efficient operating mode, but may be desirable in certain applications. The output can source or sink current in this mode. When sinking current while in forced continuous operation, current will be forced back into the main power supply potentially boosting the input supply to dangerous voltage levels-- BEWARE!
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OPERATIO
INTVCC/EXTVCC Power Power for the top and bottom MOSFET drivers and most other internal circuitry is derived from the INTVCC pin. When the EXTVCC pin is left open, an internal 5V low dropout linear regulator supplies INTVCC power. If EXTVCC is taken above 4.7V, the 5V regulator is turned off and an internal switch is turned on connecting EXTVCC to INTVCC. This allows the INTVCC power to be derived from a high efficiency external source such as the output of the regulator itself or a secondary winding, as described in the Applications Information section. Output Overvoltage Protection An overvoltage comparator, OV, guards against transient overshoots (>7.5%) as well as other more serious conditions that may overvoltage the output. In this case, the top MOSFET is turned off and the bottom MOSFET is turned on until the overvoltage condition is cleared. Power Good (PGOOD) Pin The PGOOD pin is connected to an open drain of an internal MOSFET. The MOSFET turns on and pulls the pin low when either output is not within 7.5% of the nominal output level as determined by the resistive feedback divider. When both outputs meet the 7.5% requirement, the MOSFET is turned off within 10s and the pin is allowed to be pulled up by an external resistor to a source of up to 7V. Foldback Current, Short-Circuit Detection and Short-Circuit Latchoff The RUN/SS capacitors are used initially to limit the inrush current of each switching regulator. After the controller has been started and been given adequate time to charge up the output capacitors and provide full load current, the RUN/SS capacitor is used in a short-circuit time-out circuit. If the output voltage falls to less than 70% of its nominal output voltage, the RUN/SS capacitor begins discharging on the assumption that the output is in an overcurrent and/or short-circuit condition. If the condition lasts for a long enough period as determined by the size of the RUN/SS capacitor, the controller will be shut down until the RUN/SS pin(s) voltage(s) are recycled.
U
(Refer to Functional Diagram)
This built-in latchoff can be overridden by providing a >5A pull-up at a compliance of 5V to the RUN/SS pin(s). This current shortens the soft start period but also prevents net discharge of the RUN/SS capacitor(s) during an overcurrent and/or short-circuit condition. Foldback current limiting is also activated when the output voltage falls below 70% of its nominal level whether or not the shortcircuit latchoff circuit is enabled. Even if a short is present and the short-circuit latchoff is not enabled, a safe, low output current is provided due to internal current foldback and actual power wasted is low due to the efficient nature of the current mode switching regulator. THEORY AND BENEFITS OF 2-PHASE OPERATION The LTC1628 and the LTC3728L family of dual high efficiency DC/DC controllers brings the considerable benefits of 2-phase operation to portable applications for the first time. Notebook computers, PDAs, handheld terminals and automotive electronics will all benefit from the lower input filtering requirement, reduced electromagnetic interference (EMI) and increased efficiency associated with 2-phase operation. Why the need for 2-phase operation? Up until the 2-phase family, constant-frequency dual switching regulators operated both channels in phase (i.e., single-phase operation). This means that both switches turned on at the same time, causing current pulses of up to twice the amplitude of those for one regulator to be drawn from the input capacitor and battery. These large amplitude current pulses increased the total RMS current flowing from the input capacitor, requiring the use of more expensive input capacitors and increasing both EMI and losses in the input capacitor and battery. With 2-phase operation, the two channels of the dualswitching regulator are operated 180 degrees out of phase. This effectively interleaves the current pulses drawn by the switches, greatly reducing the overlap time where they add together. The result is a significant reduction in total RMS input current, which in turn allows less expensive input capacitors to be used, reduces shielding requirements for EMI and improves real world operating efficiency.
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OPERATIO U
(Refer to Functional Diagram)
5V SWITCH 20V/DIV 3.3V SWITCH 20V/DIV INPUT CURRENT 5A/DIV INPUT VOLTAGE 500mV/DIV
IIN(MEAS) = 2.53ARMS
DC236 F03a
IIN(MEAS) = 1.55ARMS
DC236 F03b
(a) (b) Figure 3. Input Waveforms Comparing Single-Phase (a) and 2-Phase (b) Operation for Dual Switching Regulators Converting 12V to 5V and 3.3V at 3A Each. The Reduced Input Ripple with the LTC1628 2-Phase Regulator Allows Less Expensive Input Capacitors, Reduces Shielding Requirements for EMI and Improves Efficiency
Figure 3 compares the input waveforms for a representative single-phase dual switching regulator to the LTC1628 2-phase dual switching regulator. An actual measurement of the RMS input current under these conditions shows that 2-phase operation dropped the input current from 2.53ARMS to 1.55ARMS. While this is an impressive reduction in itself, remember that the power losses are proportional to IRMS2, meaning that the actual power wasted is reduced by a factor of 2.66. The reduced input ripple voltage also means less power is lost in the input power path, which could include batteries, switches, trace/connector resistances and protection circuitry. Improvements in both conducted and radiated EMI also directly accrue as a result of the reduced RMS input current and voltage. Of course, the improvement afforded by 2-phase operation is a function of the dual switching regulator's relative duty cycles which, in turn, are dependent upon the input voltage VIN (Duty Cycle = VOUT/VIN). Figure 4 shows how the RMS input current varies for single-phase and 2-phase operation for 3.3V and 5V regulators over a wide input voltage range. It can readily be seen that the advantages of 2-phase operation are not just limited to a narrow operating range, but in fact extend over a wide region. A good rule of thumb for most applications is that 2-phase operation will reduce the input capacitor requirement to that for just one channel operating at maximum current and 50% duty cycle. A final question: If 2-phase operation offers such an advantage over single-phase operation for dual switching
regulators, why hasn't it been done before? The answer is that, while simple in concept, it is hard to implement. Constant-frequency current mode switching regulators require an oscillator derived "slope compensation" signal to allow stable operation of each regulator at over 50% duty cycle. This signal is relatively easy to derive in singlephase dual switching regulators, but required the development of a new and proprietary technique to allow 2-phase operation. In addition, isolation between the two channels becomes more critical with 2-phase operation because switch transitions in one channel could potentially disrupt the operation of the other channel. These 2-phase parts are proof that these hurdles have been surmounted. They offer unique advantages for the ever-expanding number of high efficiency power supplies required in portable electronics.
3.0 2.5 SINGLE PHASE DUAL CONTROLLER
INPUT RMS CURRENT (A)
2.0 1.5 1.0 0.5 0 2-PHASE DUAL CONTROLLER
VO1 = 5V/3A VO2 = 3.3V/3A 0 10 20 30 INPUT VOLTAGE (V) 40
3728 F04
Figure 4. RMS Input Current Comparison
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APPLICATIO S I FOR ATIO
Figure 1 on the first page is a basic LTC3728L/LTC3728LX application circuit. External component selection is driven by the load requirement, and begins with the selection of RSENSE and the inductor value. Next, the power MOSFETs and D1 are selected. Finally, CIN and COUT are selected. The circuit shown in Figure 1 can be configured for operation up to an input voltage of 28V (limited by the external MOSFETs). RSENSE Selection For Output Current RSENSE is chosen based on the required output current. The current comparator has a maximum threshold of 75mV/RSENSE and an input common mode range of SGND to 1.1(INTVCC). The current comparator threshold sets the peak of the inductor current, yielding a maximum average output current IMAX equal to the peak value less half the peak-to-peak ripple current, IL. Allowing a margin for variations in the IC and external component values yields:
PLLFLTR PIN VOLTAGE (V)
50mV RSENSE = IMAX
When using the controller in very low dropout conditions, the maximum output current level will be reduced due to the internal compensation required to meet stability criterion for buck regulators operating at greater than 50% duty factor. A curve is provided to estimate this reduction in peak output current level depending upon the operating duty factor. Operating Frequency The IC uses a constant frequency phase-lockable architecture with the frequency determined by an internal capacitor. This capacitor is charged by a fixed current plus an additional current which is proportional to the voltage applied to the PLLFLTR pin. Refer to Phase-Locked Loop and Frequency Synchronization in the Applications Information section for additional information. A graph for the voltage applied to the PLLFLTR pin vs frequency is given in Figure 5. As the operating frequency
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2.5 2.0 1.5 1.0 0.5 0 200 300 400 500 OPERATING FREQUENCY (kHz) 600
3728 F05
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Figure 5. PLLFLTR Pin Voltage vs Frequency
is increased the gate charge losses will be higher, reducing efficiency (see Efficiency Considerations). The maximum switching frequency is approximately 550kHz. Inductor Value Calculation The operating frequency and inductor selection are interrelated in that higher operating frequencies allow the use of smaller inductor and capacitor values. So why would anyone ever choose to operate at lower frequencies with larger components? The answer is efficiency. A higher frequency generally results in lower efficiency because of MOSFET gate charge losses. In addition to this basic trade-off, the effect of inductor value on ripple current and low current operation must also be considered. The inductor value has a direct effect on ripple current. The inductor ripple current IL decreases with higher inductance or frequency and increases with higher VIN:
IL =
V 1 VOUT 1 - OUT ( f)(L) VIN
Accepting larger values of IL allows the use of low inductances, but results in higher output voltage ripple and greater core losses. A reasonable starting point for setting ripple current is IL=0.3(IMAX). The maximum IL occurs at the maximum input voltage.
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APPLICATIO S I FOR ATIO
The inductor value also has secondary effects. The transition to Burst Mode operation begins when the average inductor current required results in a peak current below 25% of the current limit determined by RSENSE. Lower inductor values (higher IL) will cause this to occur at lower load currents, which can cause a dip in efficiency in the upper range of low current operation. In Burst Mode operation, lower inductance values will cause the burst frequency to decrease. Inductor Core Selection Once the value for L is known, the type of inductor must be selected. High efficiency converters generally cannot afford the core loss found in low cost powdered iron cores, forcing the use of more expensive ferrite, molypermalloy, or Kool M(R) cores. Actual core loss is independent of core size for a fixed inductor value, but it is very dependent on inductance selected. As inductance increases, core losses go down. Unfortunately, increased inductance requires more turns of wire and therefore copper losses will increase. Ferrite designs have very low core loss and are preferred at high switching frequencies, so design goals can concentrate on copper loss and preventing saturation. Ferrite core material saturates "hard," which means that inductance collapses abruptly when the peak design current is exceeded. This results in an abrupt increase in inductor ripple current and consequent output voltage ripple. Do not allow the core to saturate! Molypermalloy (from Magnetics, Inc.) is a very good, low loss core material for toroids, but it is more expensive than ferrite. A reasonable compromise from the same manufacturer is Kool M. Toroids are very space efficient, especially when you can use several layers of wire. Because they generally lack a bobbin, mounting is more difficult. However, designs for surface mount are available that do not increase the height significantly. Power MOSFET and D1 Selection Two external power MOSFETs must be selected for each controller in the LTC3728L/LTC3728LX: One N-channel MOSFET for the top (main) switch, and one N-channel MOSFET for the bottom (synchronous) switch.
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The peak-to-peak drive levels are set by the INTVCC voltage. This voltage is typically 5V during start-up (see EXTVCC Pin Connection). Consequently, logic-level threshold MOSFETs must be used in most applications. The only exception is if low input voltage is expected (VIN < 5V); then, sub-logic level threshold MOSFETs (VGS(TH) < 3V) should be used. Pay close attention to the BVDSS specification for the MOSFETs as well; most of the logic level MOSFETs are limited to 30V or less. Selection criteria for the power MOSFETs include the "ON" resistance RDS(ON), Miller capacitance CMILLER, input voltage and maximum output current. Miller capacitance, CMILLER, can be approximated from the gate charge curve usually provided on the MOSFET manufacturers' data sheet. CMILLER is equal to the increase in gate charge along the horizontal axis while the curve is approximately flat divided by the specified change in VDS. This result is then multiplied by the ratio of the application applied VDS to the Gate charge curve specified VDS. When the IC is operating in continuous mode the duty cycles for the top and bottom MOSFETs are given by:
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Main Switch Duty Cycle =
VOUT VIN VIN - VOUT VIN
Synchronous Switch Duty Cycle =
The MOSFET power dissipations at maximum output current are given by:
PMAIN =
()
2 VOUT IMAX 1 + RDS(ON) + VIN 2 I VIN MAX RDR C MILLER * 2 1 1 + ( f VINTVCC - VTHMIN VTHMIN
( )( ) ( )(
)
)
PSYNC =
VIN - VOUT IMAX VIN
( ) (1+ )RDS(ON)
2
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Kool M is a registered trademark of Magnetics, Inc.
LTC3728L/LTC3728LX
APPLICATIO S I FOR ATIO
where is the temperature dependency of RDS(ON) and RDR (approximately 4) is the effective driver resistance at the MOSFET's Miller threshold voltage. VTHMIN is the typical MOSFET minimum threshold voltage. Both MOSFETs have I2R losses while the topside N-channel equation includes an additional term for transition losses, which are highest at high input voltages. For VIN < 20V the high current efficiency generally improves with larger MOSFETs, while for VIN > 20V the transition losses rapidly increase to the point that the use of a higher RDS(ON) device with lower CMILLER actually provides higher efficiency. The synchronous MOSFET losses are greatest at high input voltage when the top switch duty factor is low or during a short-circuit when the synchronous switch is on close to 100% of the period. The term (1+) is generally given for a MOSFET in the form of a normalized RDS(ON) vs Temperature curve, but = 0.005/C can be used as an approximation for low voltage MOSFETs. The Schottky diode D1 shown in Figure 1 conducts during the dead-time between the conduction of the two power MOSFETs. This prevents the body diode of the bottom MOSFET from turning on, storing charge during the deadtime and requiring a reverse recovery period that could cost as much as 3% in efficiency at high VIN. A 1A to 3A Schottky is generally a good compromise for both regions of operation due to the relatively small average current. Larger diodes result in additional transition losses due to their larger junction capacitance. CIN and COUT Selection The selection of CIN is simplified by the multiphase architecture and its impact on the worst-case RMS current drawn through the input network (battery/fuse/capacitor). It can be shown that the worst case RMS current occurs when only one controller is operating. The controller with the highest (VOUT)(IOUT) product needs to be used in the formula below to determine the maximum RMS current requirement. Increasing the output current, drawn from the other out-of-phase controller, will actually decrease the input RMS ripple current from this maximum value (see Figure 4). The out-of-phase technique typically reduces the input capacitor's RMS ripple current by a factor
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of 30% to 70% when compared to a single phase power supply solution. The type of input capacitor, value and ESR rating have efficiency effects that need to be considered in the selection process. The capacitance value chosen should be sufficient to store adequate charge to keep high peak battery currents down. 20F to 40F is usually sufficient for a 25W output supply operating at 200kHz. The ESR of the capacitor is important for capacitor power dissipation as well as overall battery efficiency. All of the power (RMS ripple current * ESR) not only heats up the capacitor but wastes power from the battery. Medium voltage (20V to 35V) ceramic, tantalum, OS-CON and switcher-rated electrolytic capacitors can be used as input capacitors, but each has drawbacks: ceramic voltage coefficients are very high and may have audible piezoelectric effects; tantalums need to be surge-rated; OS-CONs suffer from higher inductance, larger case size and limited surface-mount applicability; electrolytics' higher ESR and dryout possibility require several to be used. Multiphase systems allow the lowest amount of capacitance overall. As little as one 22F or two to three 10F ceramic capacitors are an ideal choice in a 20W to 35W power supply due to their extremely low ESR. Even though the capacitance at 20V is substantially below their rating at zero-bias, very low ESR loss makes ceramics an ideal candidate for highest efficiency battery operated systems. Also consider parallel ceramic and high quality electrolytic capacitors as an effective means of achieving ESR and bulk capacitance goals. In continuous mode, the source current of the top N-channel MOSFET is a square wave of duty cycle VOUT/VIN. To prevent large voltage transients, a low ESR input capacitor sized for the maximum RMS current of one channel must be used. The maximum RMS capacitor current is given by:
CIN RequiredIRMS IMAX
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[V (V
OUT
IN - VOUT
)]
1/ 2
VIN
This formula has a maximum at VIN = 2VOUT, where IRMS = IOUT/2. This simple worst case condition is commonly used for design because even significant deviations do not offer much relief. Note that capacitor manufacturer's
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APPLICATIO S I FOR ATIO
ripple current ratings are often based on only 2000 hours of life. This makes it advisable to further derate the capacitor, or to choose a capacitor rated at a higher temperature than required. Several capacitors may also be paralleled to meet size or height requirements in the design. Always consult the manufacturer if there is any question. The benefit of the LTC3728L/LTC3728LX multiphase clocking can be calculated by using the equation above for the higher power controller and then calculating the loss that would have resulted if both controller channels switched on at the same time. The total RMS power lost is lower when both controllers are operating due to the interleaving of current pulses through the input capacitor's ESR. This is why the input capacitor's requirement calculated above for the worst-case controller is adequate for the dual controller design. Remember that input protection fuse resistance, battery resistance and PC board trace resistance losses are also reduced due to the reduced peak currents in a multiphase system. The overall benefit of a multiphase design will only be fully realized when the source impedance of the power supply/battery is included in the efficiency testing. The drains of the two top MOSFETS should be placed within 1cm of each other and share a common CIN(s). Separating the drains and CIN may produce undesirable voltage and current resonances at VIN. The selection of COUT is driven by the required effective series resistance (ESR). Typically once the ESR requirement is satisfied the capacitance is adequate for filtering. The output ripple (VOUT) is determined by:
1 VOUT IL ESR + 8 fCOUT
Where f = operating frequency, COUT = output capacitance, and IL= ripple current in the inductor. The output ripple is highest at maximum input voltage since IL increases with input voltage. With IL = 0.3IOUT(MAX) the output ripple will typically be less than 50mV at the maximum VIN assuming: COUT Recommended ESR < 2 RSENSE and COUT > 1/(8fRSENSE)
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The first condition relates to the ripple current into the ESR of the output capacitance while the second term guarantees that the output capacitance does not significantly discharge during the operating frequency period due to ripple current. The choice of using smaller output capacitance increases the ripple voltage due to the discharging term but can be compensated for by using capacitors of very low ESR to maintain the ripple voltage at or below 50mV. The ITH pin OPTI-LOOP compensation components can be optimized to provide stable, high performance transient response regardless of the output capacitors selected. Manufacturers such as Nichicon, United Chemi-Con and Sanyo can be considered for high performance throughhole capacitors. The OS-CON semiconductor dielectric capacitor available from Sanyo has the lowest (ESR)(size) product of any aluminum electrolytic at a somewhat higher price. An additional ceramic capacitor in parallel with OS-CON capacitors is recommended to reduce the inductance effects. In surface mount applications multiple capacitors may need to be used in parallel to meet ESR, RMS current handling and load step requirements. Aluminum electrolytic, dry tantalum and special polymer capacitors are available in surface mount packages. Special polymer surface mount capacitors offer very low ESR but have lower storage capacity per unit volume than other capacitor types. These capacitors offer a very cost-effective output capacitor solution and are an ideal choice when combined with a controller having high loop bandwidth. Tantalum capacitors offer the highest capacitance density and are often used as output capacitors for switching regulators having controlled soft-start. Several excellent surge-tested choices are the AVX TPS, AVX TPSV or the KEMET T510 series of surface mount tantalums, available in case heights ranging from 2mm to 4mm. Aluminum electrolytic capacitors can be used in cost-driven applications providing that consideration is given to ripple current ratings, temperature and long term reliability. A typical application will require several to many aluminum electrolytic capacitors in parallel. A combination of the above mentioned capacitors will often result in maximizing performance and minimizing overall cost. Other capacitor types include Nichicon PL series, Panasonic SP,
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APPLICATIO S I FOR ATIO
NEC Neocap, Cornell Dubilier ESRE and Sprague 595D series. Consult manufacturers for other specific recommendations. INTVCC Regulator An internal P-channel low dropout regulator produces 5V at the INTVCC pin from the VIN supply pin. INTVCC powers the drivers and internal circuitry within the IC. The INTV CC pin regulator can supply a peak current of 50mA and must be bypassed to ground with a minimum of 4.7F tantalum, 10F special polymer, or low ESR type electrolytic capacitor. A 1F ceramic capacitor placed directly adjacent to the INTVCC and PGND IC pins is highly recommended. Good bypassing is necessary to supply the high transient currents required by the MOSFET gate drivers and to prevent interaction between channels. Higher input voltage applications in which large MOSFETs are being driven at high frequencies may cause the maximum junction temperature rating for the IC to be exceeded. The system supply current is normally dominated by the gate charge current. Additional external loading of the INTVCC and 3.3V linear regulators also needs to be taken into account for the power dissipation calculations. The total INTVCC current can be supplied by either the 5V internal linear regulator or by the EXTVCC input pin. When the voltage applied to the EXTVCC pin is less than 4.7V, all of the INTVCC current is supplied by the internal 5V linear regulator. Power dissipation for the IC in this case is highest: (VIN)(IINTVCC), and overall efficiency is lowered. The gate charge current is dependent on operating frequency as discussed in the Efficiency Considerations section. The junction temperature can be estimated by using the equations given in Note 2 of the Electrical Characteristics. For example, the IC VIN current is thermally limited to less than 67mA from a 24V supply when not using the EXTVCC pin as follows: TJ = 70C + (67mA)(24V)(34C/W) = 125C Use of the EXTVCC input pin reduces the junction temperature to: TJ = 70C + (67mA)(5V)(34C/W) = 81C The absolute maximum rating for the INTVCC Pin is 40mA. Dissipation should be calculated to also include any added
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current drawn from the internal 3.3V linear regulator. To prevent maximum junction temperature from being exceeded, the input supply current must be checked operating in continuous mode at maximum VIN. EXTVCC Connection The IC contains an internal P-channel MOSFET switch connected between the EXTVCC and INTVCC pins. When the voltage applied to EXTVCC rises above 4.7V, the internal regulator is turned off and the switch closes, connecting the EXTVCC pin to the INTVCC pin thereby supplying internal power. The switch remains closed as long as the voltage applied to EXTVCC remains above 4.5V. This allows the MOSFET driver and control power to be derived from the output during normal operation (4.7V < VOUT < 7V) and from the internal regulator when the output is out of regulation (start-up, short-circuit). If more current is required through the EXTVCC switch than is specified, an external Schottky diode can be added between the EXTVCC and INTVCC pins. Do not apply greater than 7V to the EXTVCC pin and ensure that EXTVCC < VIN. Significant efficiency gains can be realized by powering INTVCC from the output, since the VIN current resulting from the driver and control currents will be scaled by a factor of (Duty Cycle)/(Efficiency). For 5V regulators this supply means connecting the EXTVCC pin directly to VOUT. However, for 3.3V and other lower voltage regulators, additional circuitry is required to derive INTVCC power from the output. The following list summarizes the four possible connections for EXTVCC: 1. EXTVCC Left Open (or Grounded). This will cause INTVCC to be powered from the internal 5V regulator resulting in an efficiency penalty of up to 10% at high input voltages. 2. EXTVCC Connected directly to VOUT. This is the normal connection for a 5V regulator and provides the highest efficiency. 3. EXTVCC Connected to an External supply. If an external supply is available in the 5V to 7V range, it may be used to power EXTVCC providing it is compatible with the MOSFET gate drive requirements.
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4. EXTVCC Connected to an Output-Derived Boost Network. For 3.3V and other low voltage regulators, efficiency gains can still be realized by connecting EXTVCC to an output-derived voltage that has been boosted to greater than 4.7V. This can be done with either the inductive boost winding as shown in Figure 6a or the capacitive charge pump shown in Figure 6b. The charge pump has the advantage of simple magnetics. Topside MOSFET Driver Supply (CB, DB) External bootstrap capacitors CB connected to the BOOST pins supply the gate drive voltages for the topside MOSFETs. Capacitor CB in the functional diagram is charged though external diode DB from INTVCC when the SW pin is low. When one of the topside MOSFETs is to be turned on, the driver places the CB voltage across the gate-source of the desired MOSFET. This enhances the MOSFET and turns on the topside switch. The switch node voltage, SW, rises to VIN and the BOOST pin follows. With the topside MOSFET on, the boost voltage is above the input supply: VBOOST = VIN + VINTVCC. The value of the boost capacitor CB needs to be 100 times that of the total input capacitance of the topside MOSFET(s). The reverse breakdown of the external Schottky diode must be greater than VIN(MAX). When adjusting the gate drive level, the final arbiter is the total input current for the regulator. If a change is made and the input current decreases, then the efficiency has improved. If there is no change in input current, then there is no change in efficiency.
VIN OPTIONAL EXTVCC CONNECTION 5V < VSEC < 7V VIN LTC3728L/ LTC3728LX TG1 N-CH EXTVCC R6 FCB R5 SGND PGND
3728 F06a
+
CIN BAT 85 VSEC
CIN VIN
RSENSE VOUT T1 1:N
EXTVCC SW
SW
BG1 N-CH
Figure 6a. Secondary Output Loop & EXTVCC Connection
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Output Voltage The output voltages are each set by an external feedback resistive divider carefully placed across the output capacitor. The resultant feedback signal is compared with the internal precision 0.800V voltage reference by the error amplifier. The output voltage is given by the equation:
R2 VOUT = 0.8V 1 + R1
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where R1 and R2 are defined in Figure 2. SENSE+/SENSE- Pins The common mode input range of the current comparator sense pins is from 0V to (1.1)INTVCC. Continuous linear operation is guaranteed throughout this range allowing output voltage setting from 0.8V to 7.7V, depending upon the voltage applied to EXTVCC. A differential NPN input stage is biased with internal resistors from an internal 2.4V source as shown in the Functional Diagram. This requires that current either be sourced or sunk from the SENSE pins depending on the output voltage. If the output voltage is below 2.4V current will flow out of both SENSE pins to the main output. The output can be easily preloaded by the VOUT resistive divider to compensate for the current comparator's negative input bias current. The maximum current flowing out of each pair of SENSE pins is: ISENSE+ + ISENSE- = (2.4V - VOUT)/24k
VIN
+
1F
+
BAT85 0.22F BAT85
+
1F
LTC3728L/ LTC3728LX TG1 N-CH L1
VN2222LL RSENSE
BAT85 VOUT
+
COUT
BG1 N-CH PGND
+
COUT
3728 F06b
Figure 6b. Capacitive Charge Pump for EXTVCC
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APPLICATIO S I FOR ATIO
Since VOSENSE is servoed to the 0.8V reference voltage, we can choose R1 in Figure 2 to have a maximum value to absorb this current.
R1(MAX) 0.8V = 24k 2.4V - VOUT
for VOUT < 2.4V Regulating an output voltage of 1.8V, the maximum value of R1 should be 32k. Note that for an output voltage above 2.4V, R1 has no maximum value necessary to absorb the sense currents; however, R1 is still bounded by the VOSENSE feedback current. Soft-Start/Run Function The RUN/SS1 and RUN/SS2 pins are multipurpose pins that provide a soft-start function and a means to shut down the LTC3728L/LTC3728LX. Soft-start reduces the input power source's surge currents by gradually increasing the controller's current limit (proportional to VITH). This pin can also be used for power supply sequencing. An internal 1.2A current source charges up the CSS capacitor. When the voltage on RUN/SS1 (RUN/SS2) reaches 1.5V, the particular controller is permitted to start operating. As the voltage on RUN/SS increases from 1.5V to 3.0V, the internal current limit is increased from 25mV/ RSENSE to 75mV/RSENSE. The output current limit ramps up slowly, taking an additional 1.25s/F to reach full current. The output current thus ramps up slowly, reducing the starting surge current required from the input power supply. If RUN/SS has been pulled all the way to ground there is a delay before starting of approximately:
tDELAY = 1.5V C SS = 1.25s / F C SS 1.2A 3V - 1.5V C SS = 1.25s / F C SS 1.2A
(
)
tIRAMP =
(
)
By pulling both RUN/SS pins below 1V, the IC is put into low current shutdown (IQ = 20A). The RUN/SS pins can be driven directly from logic as shown in Figure 7. Diode D1 in Figure 7 reduces the start delay but allows C SS to
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VIN 3.3V OR 5V D1 RUN/SS RSS* INTVCC RSS* RUN/SS CSS CSS *OPTIONAL TO DEFEAT OVERCURRENT LATCHOFF (a) (b)
3728 F07
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Figure 7. RUN/SS Pin Interfacing
ramp up slowly providing the soft-start function. Each RUN/SS pin has an internal 6V zener clamp (See Functional Diagram). Fault Conditions: Overcurrent Latchoff The RUN/SS pins also provide the ability to latch off the controller(s) when an overcurrent condition is detected. The RUN/SS capacitor, CSS, is used initially to turn on and limit the inrush current. After the controller has been started and been given adequate time to charge up the output capacitor and provide full load current, the RUN/SS capacitor is used for a short-circuit timer. If the regulator's output voltage falls to less than 70% of its nominal value after CSS reaches 4.1V, CSS begins discharging on the assumption that the output is in an overcurrent condition. If the condition lasts for a long enough period as determined by the size of the CSS and the specified discharge current, the controller will be shut down until the RUN/SS pin voltage is recycled. If the overload occurs during startup, the time can be approximated by: tLO1 [CSS (4.1 - 1.5 + 4.1 - 3.5)]/(1.2A) = 2.7 * 106 (CSS) If the overload occurs after start-up the voltage on CSS will begin discharging from the zener clamp voltage: tLO2 [CSS (6 - 3.5)]/(1.2A) = 2.1 * 106 (CSS) This built-in overcurrent latchoff can be overridden by providing a pull-up resistor to the RUN/SS pin as shown in Figure 7. This resistance shortens the soft-start period and prevents the discharge of the RUN/SS capacitor during an over current condition. Tying this pull-up resistor to VIN, as in Figure 7a, defeats overcurrent latchoff.
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Diode-connecting this pull-up resistor to INTV CC, as in Figure 7b, eliminates any extra supply current during controller shutdown while eliminating the INTV CC loading from preventing controller start-up. Why should you defeat overcurrent latchoff? During the prototyping stage of a design, there may be a problem with noise pickup or poor layout causing the protection circuit to latch off. Defeating this feature will easily allow troubleshooting of the circuit and PC layout. The internal short-circuit and foldback current limiting still remains active, thereby protecting the power supply system from failure. After the design is complete, a decision can be made whether to enable the latchoff feature. The value of the soft-start capacitor CSS may need to be scaled with output voltage, output capacitance and load current characteristics. The minimum soft-start capacitance is given by: CSS > (COUT )(VOUT) (10 - 4) (RSENSE) The minimum recommended soft-start capacitor of CSS = 0.1F will be sufficient for most applications. Fault Conditions: Current Limit and Current Foldback The current comparators have a maximum sense voltage of 75mV resulting in a maximum MOSFET current of 75mV/RSENSE. The maximum value of current limit generally occurs with the largest VIN at the highest ambient temperature, conditions that cause the highest power dissipation in the top MOSFET. Each controller includes current foldback to help further limit load current when the output is shorted to ground. The foldback circuit is active even when the overload shutdown latch described above is overridden. If the output falls below 70% of its nominal output level, then the maximum sense voltage is progressively lowered from 75mV to 25mV. Under short-circuit conditions with very low duty cycles, the controller will begin cycle skipping in order to limit the short-circuit current. In this situation the bottom MOSFET will be dissipating most of the power but less than in normal operation. The short-circuit ripple current is determined by the minimum on-time tON(MIN) of each controller (typically 100ns), the input voltage and inductor value:
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IL(SC) = tON(MIN) (VIN/L) The resulting short-circuit current is:
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ISC =
25mV 1 - IL(SC) RSENSE 2
Fault Conditions: Overvoltage Protection (Crowbar) The overvoltage crowbar is designed to blow a system input fuse when the output voltage of the regulator rises much higher than nominal levels. The crowbar causes huge currents to flow, that blow the fuse to protect against a shorted top MOSFET if the short occurs while the controller is operating. A comparator monitors the output for overvoltage conditions. The comparator (OV) detects overvoltage faults greater than 7.5% above the nominal output voltage. When this condition is sensed, the top MOSFET is turned off and the bottom MOSFET is turned on until the overvoltage condition is cleared. The output of this comparator is only latched by the overvoltage condition itself and will therefore allow a switching regulator system having a poor PC layout to function while the design is being debugged. The bottom MOSFET remains on continuously for as long as the OV condition persists; if VOUT returns to a safe level, normal operation automatically resumes. A shorted top MOSFET will result in a high current condition which will open the system fuse. The switching regulator will regulate properly with a leaky top MOSFET by altering the duty cycle to accommodate the leakage. Phase-Locked Loop and Frequency Synchronization The IC has a phase-locked loop comprised of an internal voltage controlled oscillator and phase detector. This allows the top MOSFET turn-on to be locked to the rising edge of an external source. The frequency range of the voltage controlled oscillator is 50% around the center frequency fO. A voltage applied to the PLLFLTR pin of 1.2V corresponds to a frequency of approximately 400kHz. The nominal operating frequency range of the IC is 260kHz to 550kHz.
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APPLICATIO S I FOR ATIO
The phase detector used is an edge sensitive digital type which provides zero degrees phase shift between the external and internal oscillators. This type of phase detector will not lock up on input frequencies close to the harmonics of the VCO center frequency. The PLL hold-in range, fH, is equal to the capture range, fC: fH = fC = 0.5 fO (260kHz-550kHz) The output of the phase detector is a complementary pair of current sources charging or discharging the external filter network on the PLLFLTR pin. A simplified block diagram is shown in Figure 7. If the external frequency (fPLLIN) is greater than the oscillator frequency f0SC, current is sourced continuously, pulling up the PLLFLTR pin. When the external frequency is less than f0SC, current is sunk continuously, pulling down the PLLFLTR pin. If the external and internal frequencies are the same but exhibit a phase difference, the current sources turn on for an amount of time corresponding to the phase difference. Thus the voltage on the PLLFLTR pin is adjusted until the phase and frequency of the external and internal oscillators are identical. At this stable operating point the phase comparator output is open and the filter capacitor CLP holds the voltage. The IC's PLLIN pin must be driven from a low impedance source such as a logic gate located close to the pin. When using multiple ICs for a phase-locked system, the PLLFLTR pin of the master oscillator should be biased at a voltage that will guarantee the slave oscillator(s) ability to lock onto the master's frequency. A DC voltage of 0.7V to 1.7V applied to the master oscillator's PLLFLTR pin is recommended in order to meet this requirement. The resultant operating frequency can range from 300kHz to 500kHz. The loop filter components (CLP, RLP) smooth out the current pulses from the phase detector and provide a stable input to the voltage controlled oscillator. The filter components CLP and RLP determine how fast the loop acquires lock. Typically RLP =10k and CLP is 0.01F to 0.1F.
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Minimum On-Time Considerations Minimum on-time tON(MIN) is the smallest time duration that each controller is capable of turning on the top MOSFET. It is determined by internal timing delays and the gate charge required to turn on the top MOSFET. Low duty cycle applications may approach this minimum on-time limit and care should be taken to ensure that
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tON(MIN) <
VOUT VIN (f)
If the duty cycle falls below what can be accommodated by the minimum on-time, the controller will begin to skip cycles. The output voltage will continue to be regulated, but the ripple voltage and current will increase. The minimum on-time for each controller is approximately 100ns. However, as the peak sense voltage decreases the minimum on-time gradually increases up to about 150ns. This is of particular concern in forced continuous applications with low ripple current at light loads. If the duty cycle drops below the minimum on-time limit in this situation, a significant amount of cycle skipping can occur with correspondingly larger current and voltage ripple. FCB Pin Operation The FCB pin can be used to regulate a secondary winding or as a logic level input. Continuous operation is forced on both controllers when the FCB pin drops below 0.8V. During continuous mode, current flows continuously in the transformer primary. The secondary winding(s) draw current only when the bottom, synchronous switch is on. When primary load currents are low and/or the VIN/VOUT ratio is low, the synchronous switch may not be on for a sufficient amount of time to transfer power from the output capacitor to the secondary load. Forced continuous operation will support secondary windings providing there is sufficient synchronous switch duty factor. Thus, the FCB input pin removes the requirement that power must be drawn from the inductor primary in order to extract
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power from the auxiliary windings. With the loop in continuous mode, the auxiliary outputs may nominally be loaded without regard to the primary output load. The secondary output voltage VSEC is normally set as shown in Figure 6a by the turns ratio N of the transformer: VSEC (N + 1) VOUT However, if the controller goes into Burst Mode operation and halts switching due to a light primary load current, then VSEC will droop. An external resistive divider from VSEC to the FCB pin sets a minimum voltage VSEC(MIN):
R6 VSEC(MIN) 0.8V 1 + R5
where R5 and R6 are shown in Figure 2. If VSEC drops below this level, the FCB voltage forces temporary continuous switching operation until VSEC is again above its minimum. In order to prevent erratic operation if no external connections are made to the FCB pin, the FCB pin has a 0.18A internal current source pulling the pin high. Include this current when choosing resistor values R5 and R6. The following table summarizes the possible states available on the FCB pin:
Table 1
FCB Pin 0V to 0.75V Condition Forced Continuous Both Controllers (Current Reversal Allowed-- Burst Inhibited) Minimum Peak Current Induces Burst Mode Operation No Current Reversal Allowed Regulating a Secondary Winding Burst Mode Operation Disabled Constant Frequency Mode Enabled No Current Reversal Allowed No Minimum Peak Current
0.85V < VFCB < 4.3V
Feedback Resistors >4.8V
Voltage Positioning Voltage positioning can be used to minimize peak-to-peak output voltage excursions under worst-case transient loading conditions. The open-loop DC gain of the control
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loop is reduced depending upon the maximum load step specifications. Voltage positioning can easily be added to either or both controllers by loading the ITH pin with a resistive divider having a Thevenin equivalent voltage source equal to the midpoint operating voltage range of the error amplifier, or 1.2V (see Figure 8). The resistive load reduces the DC loop gain while maintaining the linear control range of the error amplifier. The maximum output voltage deviation can theoretically be reduced to half or alternatively the amount of output capacitance can be reduced for a particular application. A complete explanation is included in Design Solutions 10. (See www.linear-tech.com)
INTVCC RT2 ITH RT1 RC CC
3728 F08
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LTC3728L/ LTC3728LX
Figure 8. Active Voltage Positioning Applied to the LTC3728L/LTC3728LX
Efficiency Considerations The percent efficiency of a switching regulator is equal to the output power divided by the input power times 100%. It is often useful to analyze individual losses to determine what is limiting the efficiency and which change would produce the most improvement. Percent efficiency can be expressed as: %Efficiency = 100% - (L1 + L2 + L3 + ...) where L1, L2, etc. are the individual losses as a percentage of input power. Although all dissipative elements in the circuit produce losses, four main sources usually account for most of the losses in LTC3728L/LTC3728LX circuits: 1) IC VIN current (including loading on the 3.3V internal regulator), 2) INTVCC regulator current, 3) I2R losses, 4) Topside MOSFET transition losses. 1. The VIN current has two components: the first is the DC supply current given in the Electrical Characteristics table,
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APPLICATIO S I FOR ATIO
which excludes MOSFET driver and control currents; the second is the current drawn from the 3.3V linear regulator output. VIN current typically results in a small (<0.1%) loss. 2. INTVCC current is the sum of the MOSFET driver and control currents. The MOSFET driver current results from switching the gate capacitance of the power MOSFETs. Each time a MOSFET gate is switched from low to high to low again, a packet of charge dQ moves from INTVCC to ground. The resulting dQ/dt is a current out of INTVCC that is typically much larger than the control circuit current. In continuous mode, IGATECHG =f(QT+QB), where QT and QB are the gate charges of the topside and bottom side MOSFETs. Supplying INTVCC power through the EXTVCC switch input from an output-derived source will scale the VIN current required for the driver and control circuits by a factor of (Duty Cycle)/(Efficiency). For example, in a 20V to 5V application, 10mA of INTVCC current results in approximately 2.5mA of VIN current. This reduces the mid-current loss from 10% or more (if the driver was powered directly from VIN) to only a few percent. 3. I2R losses are predicted from the DC resistances of the fuse (if used), MOSFET, inductor, current sense resistor, and input and output capacitor ESR. In continuous mode the average output current flows through L and RSENSE, but is "chopped" between the topside MOSFET and the synchronous MOSFET. If the two MOSFETs have approximately the same RDS(ON), then the resistance of one MOSFET can simply be summed with the resistances of L, RSENSE and ESR to obtain I2R losses. For example, if each RDS(ON) = 30m, RL = 50m, RSENSE = 10m and RESR = 40m (sum of both input and output capacitance losses), then the total resistance is 130m. This results in losses ranging from 3% to 13% as the output current increases from 1A to 5A for a 5V output, or a 4% to 20% loss for a 3.3V output. Efficiency varies as the inverse square of VOUT for the same external components and output power level. The combined effects of increasingly lower output voltages and higher currents required by high performance digital systems is not doubling but quadrupling the importance of loss terms in the switching regulator system!
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4. Transition losses apply only to the topside MOSFET(s), and become significant only when operating at high input voltages (typically 15V or greater). Transition losses can be estimated from:
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Transition Loss = VIN
() (
I * MAX RDR * 2 1 1 + CMILLER f 5V - VTH VTH
2
()
)( )
Other "hidden" losses such as copper trace and internal battery resistances can account for an additional 5% to 10% efficiency degradation in portable systems. It is very important to include these "system" level losses during the design phase. The internal battery and fuse resistance losses can be minimized by making sure that CIN has adequate charge storage and very low ESR at the switching frequency. A 25W supply will typically require a minimum of 20F to 40F of capacitance having a maximum of 20m to 50m of ESR. The LTC3728L 2-phase architecture typically halves this input capacitance requirement over competing solutions. Other losses including Schottky conduction losses during dead-time and inductor core losses generally account for less than 2% total additional loss. Checking Transient Response The regulator loop response can be checked by looking at the load current transient response. Switching regulators take several cycles to respond to a step in DC (resistive) load current. When a load step occurs, VOUT shifts by an amount equal to ILOAD (ESR), where ESR is the effective series resistance of COUT. ILOAD also begins to charge or discharge COUT generating the feedback error signal that forces the regulator to adapt to the current change and return VOUT to its steady-state value. During this recovery time VOUT can be monitored for excessive overshoot or ringing, which would indicate a stability problem. OPTILOOP compensation allows the transient response to be optimized over a wide range of output capacitance and ESR values. The availability of the ITH pin not only allows optimization of control loop behavior but also provides a DC coupled and AC filtered closed loop response test point. The DC step, rise time and settling at this test point
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truly reflects the closed loop response. Assuming a predominantly second order system, phase margin and/or damping factor can be estimated using the percentage of overshoot seen at this pin. The bandwidth can also be estimated by examining the rise time at the pin. The ITH external components shown in the Figure 1 circuit will provide an adequate starting point for most applications.
The ITH series RC-CC filter sets the dominant pole-zero loop compensation. The values can be modified slightly (from 0.5 to 2 times their suggested values) to optimize transient response once the final PC layout is done and the particular output capacitor type and value have been determined. The output capacitors need to be selected because the various types and values determine the loop gain and phase. An output current pulse of 20% to 80% of full-load current having a rise time of 1s to 10s will produce output voltage and ITH pin waveforms that will give a sense of the overall loop stability without breaking the feedback loop. Placing a power MOSFET directly across the output capacitor and driving the gate with an appropriate signal generator is a practical way to produce a realistic load step condition. The initial output voltage step resulting from the step change in output current may not be within the bandwidth of the feedback loop, so this signal cannot be used to determine phase margin. This is why it is better to look at the ITH pin signal which is in the feedback loop and is the filtered and compensated control loop response. The gain of the loop will be increased by increasing RC and the bandwidth of the loop will be increased by decreasing CC. If RC is increased by the same factor that CC is decreased, the zero frequency will be kept the same, thereby keeping the phase shift the same in the most critical frequency range of the feedback loop. The output voltage settling behavior is related to the stability of the closed-loop system and will demonstrate the actual overall supply performance. A second, more severe transient is caused by switching in loads with large (>1F) supply bypass capacitors. The discharged bypass capacitors are effectively put in parallel with COUT, causing a rapid drop in VOUT. No regulator can alter its delivery of current quickly enough to prevent this sudden step change in output voltage if the load switch resistance is low and it is driven quickly. If the ratio of CLOAD to COUT is greater than1:50, the switch rise time
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should be controlled so that the load rise time is limited to approximately 25 * CLOAD. Thus a 10F capacitor would require a 250s rise time, limiting the charging current to about 200mA. Automotive Considerations: Plugging into the Cigarette Lighter As battery-powered devices go mobile, there is a natural interest in plugging into the cigarette lighter in order to conserve or even recharge battery packs during operation. But before you connect, be advised: you are plugging into the supply from hell. The main power line in an automobile is the source of a number of nasty potential transients, including load-dump, reverse-battery, and double-battery. Load-dump is the result of a loose battery cable. When the cable breaks connection, the field collapse in the alternator can cause a positive spike as high as 60V which takes several hundred milliseconds to decay. Reverse-battery is just what it says, while double-battery is a consequence of tow-truck operators finding that a 24V jump start cranks cold engines faster than 12V. The network shown in Figure 9 is the most straightforward approach to protect a DC/DC converter from the ravages of an automotive power line. The series diode prevents current from flowing during reverse-battery, while the transient suppressor clamps the input voltage during load-dump. Note that the transient suppressor should not conduct during double-battery operation, but must still clamp the input voltage below breakdown of the converter. Although the LTC3728L/LTC3728LX have a maximum input voltage of 30V, most applications will also be limited to 30V by the MOSFET BVDSS.
50A IPK RATING
12V
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VIN LTC3728L/ LTC3728LX
TRANSIENT VOLTAGE SUPPRESSOR GENERAL INSTRUMENT 1.5KA24A
3728 F09
Figure 9. Automotive Application Protection
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Design Example As a design example for one channel, assume VIN = 12V(nominal), VIN = 22V(max), VOUT = 1.8V, IMAX = 5A, and f = 300kHz. The inductance value is chosen first based on a 30% ripple current assumption. The highest value of ripple current occurs at the maximum input voltage. Tie the PLLFLTR pin to a resistive divider from the INTVCC pin, generating 0.7V for 300kHz operation. The minimum inductance for 30% ripple current is:
IL = VOUT VOUT 1- ( f)(L) VIN
A 4.7H inductor will produce 23% ripple current and a 3.3H will result in 33%. The peak inductor current will be the maximum DC value plus one half the ripple current, or 5.84A, for the 3.3H value. Increasing the ripple current will also help ensure that the minimum on-time of 100ns is not violated. The minimum on-time occurs at maximum VIN:
tON(MIN) =
VOUT VIN(MAX)f
=
1.8V = 273ns 22V(300kHz)
The RSENSE resistor value can be calculated by using the maximum current sense voltage specification with some accommodation for tolerances: 60mV RSENSE 0.01 5.84A Since the output voltage is below 2.4V the output resistive divider will need to be sized to not only set the output voltage but also to absorb the SENSE pin's specified input current.
0.8V R1(MAX) = 24k 2.4V - VOUT 0.8V = 24k = 32k 2.4V - 1.8V
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Choosing 1% resistors: R1 = 25.5k and R2 = 32.4k yields an output voltage of 1.816V. The power dissipation on the top side MOSFET can be easily estimated. Choosing a Fairchild FDS6982S dual MOSFET results in: RDS(ON) = 0.035/0.022, CMILLER = 215pF. At maximum input voltage with T(estimated) = 50C:
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PMAIN =
( ) [1+ (0.005)(50C - 25C)] * 2 (0.035) + (22V) 52A (4)(215pF) *
1.8V 5 22V
2
1 1 + (300kHz = 332mW 5 - 2.3 2.3
)
A short-circuit to ground will result in a folded back current of:
ISC = 25mV 1 120ns(22V) - = 2.1A 0.01 2 3.3H
with a typical value of RDS(ON) and = (0.005/C)(20) = 0.1. The resulting power dissipated in the bottom MOSFET is:
PSYNC =
22V - 1.8 V (2.1A)2 (1.125)(0.022) 22V = 100mW
which is less than under full-load conditions. CIN is chosen for an RMS current rating of at least 3A at temperature assuming only this channel is on. COUT is chosen with an ESR of 0.02 for low output ripple. The output ripple in continuous mode will be highest at the maximum input voltage. The output voltage ripple due to ESR is approximately: VORIPPLE = RESR (IL) = 0.02(1.67A) = 33mVP-P
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PC Board Layout Checklist When laying out the printed circuit board, the following checklist should be used to ensure proper operation of the IC. These items are also illustrated graphically in the layout diagram of Figure 10. The Figure 11 illustrates the current waveforms present in the various branches of the 2-phase synchronous regulators operating in the continuous mode. Check the following in your layout: 1. Are the top N-channel MOSFETs M1 and M3 located within 1cm of each other with a common drain connection at CIN? Do not attempt to split the input decoupling for the two channels as it can cause a large resonant loop.
RUN/SS1 SENSE1 + R2 R1 VOSENSE1 PLLFLTR PLLIN INTVCC SENSE1 -
PGOOD TG1 SW1
BOOST1 VIN BG1 1F CERAMIC
fIN
+
EXTVCC FCB LTC3728L/LTC3728LX INTVCC ITH1 SGND PGND BG2 BOOST2
+
3.3V
3.3VOUT ITH2 VOSENSE2
SW2 RSENSE TG2 L2 RUN/SS2 VOUT2
R3
R4
SENSE2 - SENSE2 +
Figure 10. LTC3728L/LTC3728LX Recommended Printed Circuit Layout Diagram
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2. Are the signal and power grounds kept separate? The combined IC signal ground pin and the ground return of CINTVCC must return to the combined COUT (-) terminals. The path formed by the top N-channel MOSFET, Schottky diode and the CIN capacitor should have short leads and PC trace lengths. The output capacitor (-) terminals should be connected as close as possible to the (-) terminals of the input capacitor by placing the capacitors next to each other and away from the Schottky loop described above. 3. Do the LTC3728L/LTC3728LX VOSENSE pins' resistive dividers connect to the (+) terminals of COUT? The resistive divider must be connected between the (+) terminal of
RPU PGOOD L1 RSENSE VOUT1 VPULL-UP (<7V) CB1 M1 M2 D1 COUT1 RIN CVIN VIN 1F CERAMIC M3 CB2 M4
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+
GND
CIN
+
CINTVCC
COUT2
D2
3728 F10
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CERAMIC
VIN RIN CIN
+
BOLD LINES INDICATE HIGH SWITCHING CURRENT. KEEP LINES TO A MINIMUM LENGTH.
CERAMIC
Figure 11. Branch Current Waveforms
COUT and signal ground. The R2 and R4 connections should not be along the high current input feeds from the input capacitor(s). 4. Are the SENSE - and SENSE + leads routed together with minimum PC trace spacing? The filter capacitor between SENSE + and SENSE - should be as close as possible to the IC. Ensure accurate current sensing with Kelvin connections at the SENSE resistor. 5. Is the INTVCC decoupling capacitor connected close to the IC, between the INTVCC and the power ground pins? This capacitor carries the MOSFET drivers current peaks.
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SW1 L1 RSENSE1 VOUT1 D1 COUT1
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+
RL1
SW2
L2
RSENSE2
VOUT2
D2
COUT2
+
RL2
3728 F11
An additional 1F ceramic capacitor placed immediately next to the INTVCC and PGND pins can help improve noise performance substantially. 6. Keep the switching nodes (SW1, SW2), top gate nodes (TG1, TG2), and boost nodes (BOOST1, BOOST2) away from sensitive small-signal nodes, especially from the opposites channel's voltage and current sensing feedback pins. All of these nodes have very large and fast moving signals and therefore should be kept on the "output side" of the LTC3728L/LTC3728LX and occupy minimum PC trace area.
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7. Use a modified "star ground" technique: a low impedance, large copper area central grounding point on the same side of the PC board as the input and output capacitors with tie-ins for the bottom of the INTVCC decoupling capacitor, the bottom of the voltage feedback resistive divider and the SGND pin of the IC. PC Board Layout Debugging Start with one controller on at a time. It is helpful to use a DC-50MHz current probe to monitor the current in the inductor while testing the circuit. Monitor the output switching node (SW pin) to synchronize the oscilloscope to the internal oscillator and probe the actual output voltage as well. Check for proper performance over the operating voltage and current range expected in the application. The frequency of operation should be maintained over the input voltage range down to dropout and until the output load drops below the low current operation threshold--typically 10% to 20% of the maximum designed current level in Burst Mode operation. The duty cycle percentage should be maintained from cycle to cycle in a well-designed, low noise PCB implementation. Variation in the duty cycle at a subharmonic rate can suggest noise pickup at the current or voltage sensing inputs or inadequate loop compensation. Overcompensation of the loop can be used to tame a poor PC layout if regulator bandwidth optimization is not required. Only after each controller is checked for its individual performance should both controllers be turned on at the same time. A particularly difficult region of operation is when one controller channel is nearing its current comparator trip point when the other channel is turning on its top MOSFET. This occurs around 50% duty cycle on either channel due to the phasing of the internal clocks and may cause minor duty cycle jitter.
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Short-circuit testing can be performed to verify proper overcurrent latchoff, or 5A can be provided to the RUN/ SS pin(s) by resistors from VIN to prevent the short-circuit latchoff from occurring. Reduce VIN from its nominal level to verify operation of the regulator in dropout. Check the operation of the undervoltage lockout circuit by further lowering VIN while monitoring the outputs to verify operation. Investigate whether any problems exist only at higher output currents or only at higher input voltages. If problems coincide with high input voltages and low output currents, look for capacitive coupling between the BOOST, SW, TG, and possibly BG connections and the sensitive voltage and current pins. The capacitor placed across the current sensing pins needs to be placed immediately adjacent to the pins of the IC. This capacitor helps to minimize the effects of differential noise injection due to high frequency capacitive coupling. If problems are encountered with high current output loading at lower input voltages, look for inductive coupling between CIN, Schottky and the top MOSFET components to the sensitive current and voltage sensing traces. In addition, investigate common ground path voltage pickup between these components and the SGND pin of the IC. An embarrassing problem, which can be missed in an otherwise properly working switching regulator, results when the current sensing leads are hooked up backwards. The output voltage under this improper hookup will still be maintained but the advantages of current mode control will not be realized. Compensation of the voltage loop will be much more sensitive to component selection. This behavior can be investigated by temporarily shorting out the current sensing resistor--don't worry, the regulator will still maintain control of the output voltage.
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LTC3728L/LTC3728LX
TYPICAL APPLICATIO S
59k 1M
RUN/SS1 0.1F 180pF 20k 1% 105k, 1% 1000pF SENSE1 - VOSENSE1 PLLFLTR PLLIN SENSE1 +
33pF
15k 33pF
1000pF SGND 3.3V 3.3VOUT ITH2 1000pF VOSENSE2 SENSE2 - 63.4k 1% 1000pF SENSE2 +
PGND BG2
1F 10V
+
EXTVCC FCB LTC3728L/LTC3728LX INTVCC ITH1
15k 20k 1%
180pF
0.1F
3728 F12
VIN: 7V TO 28V VOUT: 5V, 3A/3.3V, 6A/12V, 150mA SWITCHING FREQUENCY = 250kHz MI, M2: FDS6982S OR VISHAY Si4810DY L1: SUMIDA CEP123-6R3MC T1: 10H 1:1.8 -- DALE LPE6562-A262 GAPPED E-CORE OR BH ELECTRONICS #501-0657 GAPPED TOROID
Figure 12. LTC3728L/LTC3728LX High Efficiency Low Noise 5V/3A, 3.3V/5A, 12V/120mA Regulator
U
100k PGOOD TG1 SW1
VPULL-UP (<7V) PGOOD
MBRS1100T3 T1, 1:1.8 10H 0.015 M1 33F 25V
+
VOUT1 5V 3A; 4A PEAK 8 5 LT1121 ON/OFF 1 220k VOUT3 12V 120mA
0.1F BOOST1 VIN BG1
Q1
Q2 D1 3
2
10 CMDSH-3TR 0.1F
22F 50V
150F, 6.3V PANASONIC SP
+
100k
1F 25V GND
+ +
4.7F CMDSH-3TR M2 Q3 Q4
180F, 4V PANASONIC SP
VIN 7V TO 28V
BOOST2 0.1F SW2 TG2 RUN/SS2 L1 6.3H
D2
0.01
VOUT2 3.3V 5A; 6A PEAK
3728lxfa
29
LTC3728L/LTC3728LX
TYPICAL APPLICATIO S
VPULL-UP (<7V) RUN/SS1 0.1F 180pF 20k 1% 105k 1% 1000pF SENSE1 - VOSENSE1 0.01F PLLFLTR fSYNC PLLIN SW1 0.1F BOOST1 VIN BG1 10 FCB EXTVCC LTC3728L/LTC3728LX ITH1 INTVCC 1500pF SGND 3.3V 3.3VOUT ITH2 1000pF VOSENSE2 SENSE2 180pF 63.4k 1% 1000pF SENSE2 + RUN/SS2
-
10k 100pF
1000pF
+
8.06k 100pF
4.75k 20k 1%
0.1F VIN: 7V TO 28V VOUT: 5V, 4A/3.3V, 5A SWITCHING FREQUENCY = 250kHz TO 550kHz M1, M2: FDS6982S OR VISHAY Si4810DY L1, L2: SUMIDA CDEP105-4R3MC-88 OUTPUT CAPACITORS: PANASONIC SP SERIES
Figure 13. LTC3728L/LTC3728LX 5V/4A, 3.3V/5A Regulator with External Frequency Synchronization
30
U
PGOOD TG1
PGOOD
L1 4.3H
SENSE1 +
0.008
VOUT1 5V/4A
Q1 PIN 4 M1 1F 50V
Q2
+
CMDSH-3TR 0.1F 1F
22F 50V
150F, 6.3V
GND
+
4.7F, 10V CMDSH-3TR
1F 50V
PGND BG2 BOOST2 0.1F SW2 TG2 L2 4.3H M2 PIN 4 Q3 Q4
180F, 4V
VIN 7V TO 28V
0.008
VOUT2 3.3V/5A
3728 F13
3728lxfa
LTC3728L/LTC3728LX
PACKAGE DESCRIPTIO U
(For purposes of clarity, drawings are not to scale) GN Package 28-Lead Plastic SSOP (Narrow .150 Inch)
(Reference LTC DWG # 05-08-1641)
.045 .005 .386 - .393* (9.804 - 9.982) 28 27 26 25 24 23 22 21 20 19 18 17 1615 .033 (0.838) REF
.254 MIN
.150 - .165 .229 - .244 (5.817 - 6.198) .150 - .157** (3.810 - 3.988)
.0165 .0015 RECOMMENDED SOLDER PAD LAYOUT
.0250 TYP 1 .053 - .069 (1.351 - 1.748) 23 4 56 7 8 9 10 11 12 13 14 .004 - .009 (0.102 - 0.249)
.015 .004 x 45 (0.38 0.10) .0075 - .0098 (0.191 - 0.249) 0 - 8 TYP
.016 - .050 (0.406 - 1.270) NOTE: 1. CONTROLLING DIMENSION: INCHES INCHES 2. DIMENSIONS ARE IN (MILLIMETERS)
.008 - .012 (0.203 - 0.305)
.0250 (0.635) BSC
3. DRAWING NOT TO SCALE *DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE **DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
GN28 (SSOP) 0502
UH Package 32-Lead Plastic QFN (5mm x 5mm)
(Reference LTC DWG # 05-08-1693)
0.57 0.05 5.00 0.10 (4 SIDES) 0.75 0.05 0.00 - 0.05 BOTTOM VIEW--EXPOSED PAD R = 0.115 TYP 31 32 0.40 0.10
PIN 1 TOP MARK 5.35 0.05 4.20 0.05 3.45 0.05 (4 SIDES)
1 2
3.45 0.10 (4-SIDES)
(UH) QFN 0102
0.23 0.05 PACKAGE OUTLINE 0.50 BSC RECOMMENDED SOLDER PAD LAYOUT
0.200 REF NOTE: 1. DRAWING PROPOSED TO BE A JEDEC PACKAGE OUTLINE M0-220 VARIATION WHHD-(X) (TO BE APPROVED) 2. DRAWING NOT TO SCALE 3. ALL DIMENSIONS ARE IN MILLIMETERS 4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.20mm ON ANY SIDE 5. EXPOSED PAD SHALL BE SOLDER PLATED
0.23 0.05 0.50 BSC
3728lxfa
Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
31
LTC3728L/LTC3728LX
TYPICAL APPLICATIO U
IIN 12VIN I1 OPEN PHASMD TG1 0 180 BUCK: 2.5V/15A I2 BUCK: 2.5V/15A 2.5VO/30A I1 I2 I3 BUCK: 1.5V/15A BUCK: 1.8V/15A I4
3728 F14
CIN IIN*
U1 TG2 LTC3729 90 CLKOUT
I3 TG1 U2 TG2 LTC3728L/ LTC3728LX 90 PLLIN 90 270 1.5VO/15A 1.8VO/15A
I4 *INPUT RIPPLE CURRENT CANCELLATION INCREASES THE RIPPLE FREQUENCY AND REDUCES THE RMS INPUT RIPPLE CURRENT THUS, SAVING INPUT CAPACITORS
Figure 14. Multioutput PolyPhase Application
RELATED PARTS
PART NUMBER LTC1628/LTC1628-PG/ LTC1628-SYNC LTC1629/ LTC1629-PG LTC1702A LTC1708-PG LT1709/ LT1709-8 LTC1735 LTC1736 LTC1778/LTC1778-1 LTC1929/ LTC1929-PG LTC3708 LTC3711 LTC3728 LTC3729 LTC3731 DESCRIPTION 2-Phase, Dual Output Synchronous Step-Down DC/DC Controller 20A to 200A PolyPhaseTM Synchronous Controllers No RSENSE 2-Phase Dual Synchronous Step-Down Controller 2-Phase, Dual Synchronous Controller with Mobile VID High Efficiency, 2-Phase Synchronous Step-Down Switching Regulators with 5-Bit VID High Efficiency Synchronous Step-Down Switching Regulator High Efficiency Synchronous Controller with 5-Bit Mobile VID Control No RSENSE Current Mode Synchronous Step-Down Controllers 2-Phase Synchronous Controllers Dual, 2-Phase, DC/DC Controller with Output Tracking No RSENSE Current Mode Synchronous Step-Down Controller with Digital 5-Bit Interface Dual, 550kHz, 2-Phase Synchronous Step-Down Controller 20A to 200A, 550kHz PolyPhase Synchronous Controller 3- to 12-Phase Step-Down Synchronous Controller COMMENTS Reduces CIN and COUT, Power Good Output Signal, Synchronizable, 3.5V VIN 36V, IOUT up to 20A, 0.8V VOUT 5V Expandable from 2-Phase to 12-Phase, Uses All Surface Mount Components, No Heat Sink, VIN up to 36V 550kHz, No Sense Resistor 3.5V VIN 36V, VID Sets VOUT1, PGOOD 1.3V VOUT 3.5V, Current Mode Ensures Accurate Current Sharing, 3.5V VIN 36V Output Fault Protection, 16-Pin SSOP Output Fault Protection, 24-Pin SSOP, 3.5V VIN 36V Up to 97% Efficiency, 4V VIN 36V, 0.8V VOUT (0.9)(VIN), IOUT up to 20A Up to 42A, Uses All Surface Mount Components, No Heat Sinks, 3.5V VIN 36V Current Mode, No RSENSE, Up/Down Tracking, Synchronizable Up to 97% Efficiency, Ideal for Pentium(R) III Processors, 0.925V VOUT 2V, 4V VIN 36V, IOUT up to 20A Dual 180 Phased Controllers, VIN 3.5V to 35V, 99% Duty Cycle, 5x5QFN, SSOP-28 Expandable from 2-Phase to 12-Phase, Uses all Surface Mount Components, VIN up to 36V 60A to 240A Output Current, 0.6V VOUT 6V, 4.5V VIN 32V
No RSENSE and PolyPhase are trademarks of Linear Technology Corporation. Pentium is a registered trademark of Intel Corporation.
3728lxfa
32 Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 q FAX: (408) 434-0507
q
LT/TP 0104 1K REV A * PRINTED IN USA
www.linear.com
(c) LINEAR TECHNOLOGY CORPORATION 2002

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